1. Field of the Invention
This invention relates to switching converters particularly adapted to switch at relatively high frequencies and, in particular, to such converters that achieves switching on and off under zero-voltage conditions, whereby switching losses associated with semiconductor switching devices may be substantially eliminated.
2. Cross-Reference To Copending Applications
Attention is drawn to the following copending, commonly assigned applications, all/each incorporated specifically by reference into the instant specification:
(1) "RESONANT CONVERTERS WITH SECONDARY-SIDE RESONANCE", filed Apr. 28, 1986 in the names of Fred C. Lee and Kwang-Hwa Liu, Ser. No. 856,775; and
(2) "ZERO-CURRENT SWITCHING CONVERTERS", filed on even date in the names of Fred C. Lee and Kwang-Haw Liu, Ser. No. 877,184 filed June 20, 1986.
3. Description of the Prior Art
Electronic power processing technology has evolved around two fundamentally different circuit schemes: (1) duty-cycle modulation, commonly known as Pulse Width Modulation (PWM), and resonance. The PWM technique processes power by interrupting the power flow and controlling the duty cycle, thus, resulting in pulsating current and voltage waveforms. The resonant technique processes power in a sinusoidal form. Due to circuit simplicity and ease of control, the PWM technique has been used predominantly in today's power electronics industries, particularly, in low-power power supply applications, and is quickly becoming a mature technology. Resonant technology, although well established in high-power SCR motor drives and uninterrupted power supplies, has not been widely used in low-power dc/dc converter applications due to its circuit complexity.
In conventional PWM converters, a switching device typically in the form of an available semiconductor switch turns on and off repetitively at a rate typically in the range of 30-50 kH.sub.z and at high current levels to achieve power conversion and output voltage regulation. Such converters employ magnetic and capacitive components for energy storage/transfer and ripple filtering. With the advent of the power MOSFETs, switching speed may be increased as high as tens of MHz. Operating such magnetic and capacitive components at high frequencies reduces their size and cost. In typical PWM switching converters, the impedance of such reactive components is coupled in circuit with the semiconductor switches. As the switching frequency is increased, such reactive components adversely affects these switches. As the switch is turned on and off rapidly, switching transients involving high levels of current and voltage occur, whereby high switching stresses and losses are imposed upon the semiconductor switch. When such a switch is switched or "forced off", the energy still present in the coupled inductive element imposes high current and high voltage and, thus, high switching stress and loss on the switch. Furthermore, the pulsating current waveforms resulting from rapid switching, cause severe electromagnetic interation (EMI) problems as the switching frequency is increased. Nonetheless, it is desired to switch such semiconductor switches at higher switching frequencies to improve the dynamic and response time of the voltage control and regulation and, at the same time, minimize the size and cost of the inductive and capacitive elements employed in such converters. However, as the switching frequency increases, the above-noted switching stresses and losses increase and the converter's overall efficiency and reliability decrease.
To reduce switching stress and loss, the technique of "zero-current switching" has been described in "Resonant Switching Power Conversion Technique," by E. E. Buchanan and E. J. Miller, IEEE Power Electronics Specialists Conference, 1975 Record, pp. 188-193 and in "Resonant Switching Power Conversions," by E. J. Miller, IEEE Power Electronics Specialists Conferences, 1976 Record, pp. 206-211. Such "zero-current switching" technique utilizes an LC resonant tank circuit to force the current through the semiconductor switch to oscillate, whereby the semiconductor switch turns off at zero current level, thereby drastically reducing switching stresses and losses.
To generalize the zero-current switching technique, the concept of a resonant switch was described in "Resonant Switches--A Unified Approach to Improve Performance of Switching Converters," by the inventors of this invention, IEEE International Telecommunications Energy Conference, 1984 Proceedings, pp. 344-351. This paper described the use of "resonant switches" in various conventional pulse-width modulated switching converters to achieve "zero-current switching". Generally, such resonant switches are a subcircuit comprising a semiconductor switch S.sub.1, a resonance inductor L.sub.r, and a resonance capacitor C.sub.r. There are two types of resonant switch configurations as shown respectively in FIGS. 1A and 1B, an L-type and an M-type resonant switch. In both cases, the inductor L.sub.r is connected in series with the switch S.sub.1 to slow down the current change rate, and the capacitor C.sub.r is added as an auxiliary energy storage/transfer element. If switch S.sub.1 is a device without reverse voltage blocking capability or contains an internal anti-parallel diode, an additional diode D.sub.1 is needed and should be connected in series with the switch S.sub.1 and the inductor L.sub.r. The inductor L.sub.r and the capacitor C.sub.r together constitute a series resonant circuit with respect to the switch S.sub.1. When the switch S.sub.1 conducts, current flows through switch S.sub.1 and inductor L.sub.r into the capacitor C.sub.r with a quasi-sinusoidal waveform. As the inductor current drops to zero, the capacitor voltage V.sub.c is charged up with a negative polarity with respect to switch S.sub.1, thus commutating off the switch S.sub.1. The resonant switch therefore, provides zero-current switching properties during both turn on and turn off.
A conventional buck converter is illustrated in FIG. 2A, as comprising a switch S.sub.1 for applying upon being rendered conductive, a voltage source V.sub.s across a free-wheeling diode D. The free-wheeling diode D is coupled to a filter circuit comprised of an output inductor L.sub.o disposed in circuit with an output capacitor C.sub.o which is connected in parallel with an output resistor R.sub.o. This conventional buck converter is modified as shown in FIG. 2B by the addition of the L-type resonant switch, as first shown in FIG. 1A, between voltage source V.sub.s and the free-wheeling diode D. The output inductance L.sub.o is selected to be much larger than inductance L.sub.r, thus making the resonant frequency of the filter circuit comprised of capacitor C.sub.o and the inductor L.sub.o much smaller than that of the resonant circuit comprised of the capacitor of C.sub.r and the resonant inductor L.sub.r. It is also assumed that inductor L.sub.o is sufficiently large so that the current I.sub.2 through the inductor L.sub.o, remains relatively constant throughout a switching cycle.
The operation of the buck quasi-resonant converter employing the L-type resonant switch as shown in FIG. 2B, will now be explained with reference to the waveforms as shown in FIGS. 3A to 3D. Before time T.sub.0, the semiconductor switch S.sub.1 is turned off, whereby the free-wheeling diode D carries the output current I.sub.o with the capacitor voltage V.sub.c clamped at zero. In the first of four distinct stages, the semiconductor switch S.sub.1 is turned on at time T.sub.0, whereby the input current I.sub.1 flowing through the semiconductor switch S.sub.1 and the resonant inductor L.sub.r rises linearly as shown in the waveform of FIG. 3B between times T.sub.0 and T.sub.1. Between times T.sub.0 and T.sub.1, the output current I.sub.2 shifts gradually from the path through the free-wheeling diode D to the path through the semiconductor switch S.sub.1 and the resonant inductor L.sub.r.
At time T.sub.1, the current I.sub.1 becomes equal to current I.sub.2, whereby the free-wheeling diode D is turned off and, as seen in FIG. 3B, the current I.sub.1 begins to charge capacitor C.sub.r. As seen in FIG. 3B, the flow of the current of I.sub.1 through the resonant inductor L.sub.r and the voltage V.sub.c appearing on resonant capacitor C.sub.r is substantially sinusoidal rising to a peak and falling back to zero at time T.sub.2. As shown in FIG. 3D, the voltage V.sub.c rises to a peak of approximately 2V.sub.s shortly before time T.sub.2, whereby a reverse voltage of V.sub.c -V.sub.s is applied to the semiconductor switch S.sub.1 commutating it off naturally at time T.sub.2. As shown in FIG. 3B, zero current is flowing in the semiconductor switch S.sub.1 at time T.sub.2, when it is commutated off. As shown in FIG. 3D, the capacitor C.sub.r discharges in the time interval from time T.sub.2 to time T.sub. 3. The capacitor voltage V.sub.c drops linearly to zero at time T.sub.3. In the fourth stage from time T.sub.3 to time T.sub.4, the output current I.sub.2 flows through the free-wheeling diode D and, with the switch S.sub.1 open, the resonant capacitor C.sub.r is clamped to zero voltage. At time T.sub.4, the switch S.sub.1 turns on again, starting the next switching cycle.
FIG. 2C shows a buck quasi-resonant converter circuit in which the resonant capacitor C.sub.r is coupled in parallel between the voltage source V.sub.s and the resonant inductor L.sub.r instead of in parallel with the free-wheeling diode D, whereby an M-type resonant switch, as shown first in FIG. 1B, is formed. The modified buck quasi-resonant converter of FIG. 2C operates in four stages in a manner similar to the operation of the buck quasi-resonant converter as described above with respect to FIG. 2B.
As shown in FIGS. 2B and 2C, such zero-current quasi-resonant converters utilize the principal of inductive or capacitive energy storage and transfer in a similar fashion as PWM converters. Further, the zero-current quasi-resonant converters of FIGS. 2B and 2C employ an LC tank circuit comprised of the resonant inductor L.sub.r and the resonant capacitor C.sub.r. The LC tank circuit is coupled close to the power switch S.sub.1 and is used not only to shape the current and voltage waveforms of the switch S.sub.1, but also serves as an intermediate energy tank that stores and transfers energy from the voltage source V.sub.s to the output LC tank circuit. The switch S.sub.1 takes the form of such power processing semiconductor switching devices as bipolar transistors and MOSFETs. The conduction loss of bipolar transistor and MOSFETs remains relatively constant at high switching frequencies, while the switching losses often increase in direct proportion to the switching frequency.
The switching losses can be categorized in the following two forms: (1) the turn off loss and (2) the turn on loss. A minority-carrier device such as a bipolar transistor has a relatively longer turn off storage time and (current) fall time. For PWM converters, the voltage V.sub.ce across the transistor's output terminals at turn off, increases rapidly to a high level before the current I.sub.c starts to drop. The simultaneous presence of high voltage and high current levels causes a certain amount of energy to be dissipated within such devices. In contrast with a bipolar transistor, a majority carrier device such as a can switch at much higher switching frequencies with no storage time and much shorter fall time, whereby there is a relatively small turn off loss.
Switching devices of conventional PWM DC-to-DC converters, as well as the semiconductor switches S.sub.1 as employed in the quasi-resonant converters of FIGS. 2B and 2C, turn on at relatively high voltage levels. As seen in FIG. 3C, the voltage V.sub.ds imposed upon the switch S.sub.1 at turn on time T.sub.0, is relatively high. Since bipolar transistors and MOSFETs have respectively intrinsic parasitic capacitances C.sub.ce and C.sub.ds at their output terminals, turn on at high voltage levels induces a turn on loss of 1/2 CV.sup.2 F.sub.s, where C is the value of the parasitic drain-to-source or the collector-to-emitter capacitance, V is the voltage across the output terminals of the switch S.sub.1 before turn on, and F.sub.s is the switching frequency. This turn on loss becomes significant as the switching frequency is increased. As shown in FIGS. 2B and 2C, the switching device S.sub.1 is associated with the resonant inductor L.sub.r and the resonant capacitor C.sub.r, which serve to store and transfer energy. However, when the switch S.sub.1 is turned on at a high voltage level, this energy is stored or trapped within the parasitic capacitance of the switch S.sub.1, resulting in the afore-ascribed turn on loss. Thus, it is seen that the zero-current quasi-resonant converters as shown in FIGS. 2B and 2C cannot alleviate the problem of high switching losses associated with the inherent, parasitic capacitance of such power switches as bipolar transistors and MOSFETs.
A MOSFET has an insulated gate structure, which results in relatively high gate-to-drain capacitance C.sub.GD and gate-to-source capacitance C.sub.GS. Along with the junction depletion-capacitance C.sub.DS, these capacitances play important roles in the switching behavior of MOSFETs. In a typical resistive-load switching circuit as shown in FIGS. 2D and 2E, the gate voltage is switched between V.sub.G1 and V.sub.G2 and the drain voltage is switched between 0 and V.sub.D. Before the turn on of the MOSFET Q as shown in FIG. 2D, the voltage states on capacitances C.sub.GD, C.sub.GS and C.sub.DS are (V.sub.D +V.sub.G2),-V.sub.G2 and V.sub.D, respectively. After the MOSFET Q is completely turned on by the gate drive, the voltage states of the capacitances become respectively -V.sub.G1, V.sub.G1 and 0, as shown in FIG. 2E. The voltage changes are -(V.sub.D +V.sub.G1 +V.sub.G2), (V.sub.G1 +V.sub.G2), and -V.sub.D. For the gate drive circuit, it must supply charging currents to capacitance C.sub.GD as well as to capacitance C.sub.GS, as the voltages across these devices change abruptly. This phenomenon of displacement current due to high dv/dt across the gate capacitance C.sub.GD is similar to the Miller effect occurring in amplifier circuits, although in this case it is of a large-signal nature. More specifically, the Miller effect transfers the rate of change of the voltage V.sub.DS across the output of the MOSFET Q to its gate through the capacitance C.sub.GD, whereby if the MOSFET Q turns on when a voltage is applied to its output, a resulting ringing voltage will be induced to its gate. Potentially, the ringing voltage applied to the MOSFET's gate may render the normal operation of the MOSFET impossible.